Circuit arrangement

ABSTRACT

In an up-converter supplied by the mains supply and operating in the critical mode, the on-time of the switching element ( 2 ) is modulated so that it is increased in the vicinity of the zero crossings of the mains supply voltage. As a result the THD is lowered.

The invention relates to a circuit arrangement comprising:

-   -   circuit input terminals for connection to a low frequency supply        voltage source,        -   circuit output terminals,        -   a rectifier coupled to the circuit input terminals and            equipped with rectifier output terminals,        -   a DC-DC-converter coupled between the rectifier output            terminals and the circuit output terminals and comprising            -   an inductive element,            -   a unidirectional element coupled to the inductive                element,            -   a switching element coupled to the inductive element and                the unidirectional element,            -   a control circuit coupled to a control electrode of the                switching element for rendering the switching element                alternately conductive and non-conductive at a high                frequency and for generating a control signal for                controlling the conduction time of the switching element                in each high frequency period, said control circuit                comprising a circuit part I for generating a first                signal S1 that is either a constant or exclusively                depends on the actual voltage between the output                terminals and a desired value of the voltage between the                output terminals.

The invention also relates to a ballast circuit for operating a lamp.

A circuit arrangement as mentioned in the opening paragraph is knownfrom Texas Instruments application note U-132 for the integrated circuitUC 3852. The DC-DC-converter comprised in the known circuit arrangementis an up-converter that is supplied by a rectified low frequency supplyvoltage and is operated in the critical mode. This means that theswitching element is rendered conductive when the current through theinductive element has dropped to zero. This has the advantage that thepower dissipation in the unidirectional element generally indicated asreverse recovery loss is very limited, so that operation in the criticalmode is relatively efficient. The current through the inductive elementhas the shape of a series of triangles. Normally it is desirable thatthe average value (averaged over a high frequency period) of the currentthrough the inductive element is proportional to the rectified lowfrequency supply voltage to obtain a high power factor and low THD. Oneway of achieving this is to maintain the conduction time of theswitching element at the same constant value during half a period of thelow frequency supply voltage: in the known circuit arrangement thecontrol signal is identical to the signal S1 and the signal S1 is onlyinfluenced by the actual and the desired value of the voltage betweenthe output terminals. In other words the shape of the current throughthe inductive element is controlled by means of open loop control orfeed forward control. Circuit part I increases the conduction time incase the amplitude of the low frequency supply voltage decreases or incase the power consumed from the output terminals increases, in order tomaintain the voltage between the output terminals at a value thatsubstantially equals the desired value and is independent from theamplitude of the low frequency supply voltage or the amount of powerconsumed at the output terminals. Similarly, circuit part I decreasesthe conduction time in case the amplitude of the low frequency supplyvoltage increases or in case the power consumed from the outputterminals decreases.

Alternatively circuit part I will also increase or decrease theconduction time of the switching element in case a user of the circuitarrangement demands a higher voltage or a lower voltage respectivelybetween the output terminals and therefore adjusts the desired value ofthe voltage between the output terminals. However, during stationaryoperating conditions, the changes in the conduction time are takingplace at a comparatively slow rate so that the signal S1 and thereforethe conduction time of the switching element are substantially constantover half a period of the low frequency supply voltage. Although such asubstantially constant value of the conduction time of the switchingelement should theoretically result in a very high power factor and alow THD, there are some deviations in practical circuits. Since there isalways some input capacitance present at the input of theDC-DC-converter, such as a filter capacitor or a parasitary capacitance,the supply voltage of the DC-DC-converter differs from the rectifiedmains since in the vicinity of the zero crossings of the low frequencysupply voltage the voltage across the input capacitance will notcompletely drop to zero. As a consequence the amplitude of the lowfrequency supply voltage is lower than the voltage across the inputcapacitance so that no current is drawn from the low frequency supplyvoltage source in the vicinity of the zero crossings of the lowfrequency supply voltage. The resulting current distortion lowers thepower factor and increases the THD.

It is remarked that the same problem can arise in a circuit arrangementin which the signal S1 is a constant that is not influenced by anyoperational parameter. In this latter case the constant value of signalS1 will cause the power factor to be comparatively high and the THD tobe comparatively low, but the voltage between the output terminals willchange when the maximum amplitude of the low frequency supply voltage orthe power consumed at the output terminals changes.

Another problem occurring in the vicinity of the zero crossings of thelow frequency supply voltage is the following. Because of the lowamplitude of the low frequency supply voltage, the amount of energy thatis stored in the inductive element during each high frequency period isvery small. When the switching element is rendered non-conductive thisenergy should cause a current to flow through the unidirectional elementto the circuit output terminals. However, before the unidirectionalelement becomes conductive, the energy stored in the inductive elementfirst charges the parasitary capacitance associated with a commonterminal of the inductive element, the unidirectional element and theswitching element. In case the energy stored in the inductive element isso low that it does not suffice to charge the parasitary capacitance toa voltage high enough to render the unidirectional element conductive,no net current flows from the low frequency supply voltage source to thecircuit output terminals. Also this latter problem causes the currentdrawn from the low frequency supply voltage to become zero in thevicinity of the zero crossings of the low frequency supply voltage sothat the power factor is lowered and THD is increased.

The invention aims to provide a circuit arrangement wherein the controlcircuit is comparatively simple and therefore cheap while the controlcircuit yet controls the switching element in such a way that acomparatively high power factor is obtained while the THD iscomparatively low.

A circuit arrangement as mentioned in the opening paragraph is thereforeaccording to the invention characterized in that the control circuit isfurther equipped with a circuit part II for generating a second signalS2 that is periodical and has the same phase and frequency as therectified low frequency supply voltage and for generating the controlsignal by superimposing the second signal S2 on the first signal S1.

The control signal thus is the sum of signal S1 and signal S2. Since thesecond signal S2 varies over half a period of the supply voltage thesame is true for the control signal, so that the on-time of theswitching element varies over each half period of the low frequencysupply voltage. The shape of signal S2 is chosen so that the on-time ofthe switching element is increased in the vicinity of the zero crossingsof the low frequency supply voltage. As a result, more current is drawnfrom the low frequency supply voltage source in the vicinity of the zerocrossings of the low frequency supply voltage so that the power factoris increased and the THD is lowered. It be mentioned that also in caseof a circuit arrangement according to the invention the current drawnfrom the mains supply is zero as long as the momentary amplitude of themains supply voltage is lower than the voltage across the inputcapacitance. However, as soon as the momentary amplitude of the mainssupply is higher than the voltage across the input capacitance, thecircuit part II ensures that a much higher current is drawn from themains supply than is the case in prior art circuits in which theconduction time is maintained at a constant value during each half aperiod of the mains supply voltage. It is this increased mains currentin the vicinity of the zero crossings of the mains supply voltage, whenthe mains voltage amplitude is higher than the voltage across the inputcapacitance, that improves the power factor and lowers the THD.

In the known circuit arrangement in which the conduction time of theswitching element is controlled at a substantially constant value duringeach half period of the low frequency supply voltage, the frequency atwhich the switching element is controlled reaches the highest value inthe vicinity of the zero crossings of the low frequency supply voltage.In a circuit arrangement according to the invention, the circuit partII, by increasing the conduction time, lowers this highest frequency. Asa result the frequency range within which the control circuit mustfunction is decreased, simplifying the design of the control circuit.

Good results have been obtained for embodiments of a circuit arrangementaccording to the invention, wherein the second signal is proportional tothe rectified low frequency supply voltage. It is remarked that the signof the proportionality factor depends on the relation between controlsignal and conduction time. In case the circuit arrangement is sodesigned that a higher value of the control signal corresponds to ashorter conduction time the proportionality factor is positive so thatthe signal S2 is directly proportional to the rectified low frequencysupply voltage. However, in case the circuit arrangement is so designedthat a higher value of the control signal corresponds to a longerconduction time, the proportionality factor is negative so that thesignal S2 is in fact invertedly proportional to the rectified lowfrequency supply voltage. Since the rectified low frequency supplyvoltage is present at the output of the rectifier, the circuit part IIcan be comparatively simple. The circuit part II may for instancecomprise a coupling capacitor coupled between the rectifier and circuitpart I or it may comprise a series arrangement of a first impedance anda second impedance coupled between the rectifier output terminals andwherein a common terminal of the two impedances is coupled to an outputof the circuit part I.

Even better results have been obtained for embodiments of a circuitarrangement according to the invention, wherein the second signal isproportional to the rectified low frequency supply voltage when theamplitude of the rectified low frequency supply voltage is lower than apredetermined reference value and substantially equal to thepredetermined reference value when the amplitude of the rectified lowfrequency supply voltage is higher than the predetermined referencevalue. It has been found that the resulting modulation of the on-time ofthe switching element over half a period of the low frequency supplyvoltage increased the power factor and lowered the THD even further withrespect to embodiments in which the signal S2 is proportional to therectified low frequency supply voltage. Such a shape of the signal S2can be realized in a comparatively simple way, in case the circuit partII comprises a zener diode. The circuit part I may comprise a seriesarrangement of a first impedance and a second impedance coupled betweenthe rectifier output terminals and wherein a common terminal of the twoimpedances is coupled to an output of the circuit part I, and wherein atleast one of the impedances is a voltage dependant impedance. The firstimpedance may comprise an ohmic resistor while the second impedancecomprises a parallel arrangement of an ohmic resistor and a zener diode.

Good results have also been obtained for embodiments wherein the firstimpedance comprises a series arrangement of a zener diode and an ohmicresistor and the second impedance comprises an ohmic resistor, and forembodiments wherein the first impedance comprises an ohmic resistor andthe second impedance comprises an ohmic resistor shunted by a diode.Good results have further been obtained for embodiments in which thefirst impedance comprises a series arrangement of a zener diode and anohmic resistor, while the second impedance comprises an ohmic resistorshunted by a diode.

Very low values of the THD have been obtained for embodiments of acircuit arrangement according to the invention, wherein theDC-DC-converter is an up-converter. Because of its low THD and highpower factor combined with a very simple design of the control circuit,a circuit arrangement according to the invention is very suitable foruse in a ballast circuit.

Embodiments of a circuit arrangement according to the invention will bedesribed making use of a drawing. In the drawing

FIG. 1 shows an embodiment of a circuit arragement according to theinvention with a load connected to it, and

FIGS. 2-7 show different implementations of a part of the circuitarrangement shown in FIG. 1.

In FIGS. 1, 4 and 5 are circuit input terminals for connection to a lowfrequency supply voltage source. Circuit input terminals 4 and 5 areconnected to respective input terminals of diode bridge 7, forming arectifier. 8 and 9 are rectifier output terminals of the rectifierformed by the diode bridge 7. Diode bridge output terminals 8 and 9 areconnected by means of a series arrangement of inductive element 10 andswitching element 12. Switching element 12 is shunted by a seriesarrangement of a diode 11, forming a unidirectional element, and acapacitor 15. A common terminal 16 of diode 11 and capacitor 15 forms afirst circuit output terminal of the circuit arrangement. A secondcircuit output terminal 17 of the circuit arrangement is situatedbetween diode bridge output terminal 9 and capacitor 15. A load isconnected between output terminals 16 and 17. An output terminal ofcircuit part 13 is connected to a control electrode of switching element12. Circuit part 13 is a circuit part for rendering switching element 12alternately conductive and non-conductive at a high frequency. An inputterminal of circuit part 13 is connected to an output terminal ofcircuit part I. Circuit part I is a circuit part for generating a signalS1 that in this embodiment exclusively depends on a desired value andthe actual value of the voltage between the output terminals. An outputterminal of circuit part II is coupled to circuit part I. In FIG. 1 thiscoupling is indicated by means of a dotted line. Circuit part II is acircuit part for generating a second signal S2 that is periodical andhas the same phase and frequency as the rectified low frequency supplyvoltage and for generating the control signal by superimposing thesecond signal S2 on the first signal S1. Respective input terminals ofcircuit part II are connected to the diode bridge output terminals 8 and9 respectively. Circuit parts 13, I and II together form a controlcircuit coupled to the control electrode of the switching element.Inductive element 10, switching element 12, diode 11 and the controlcircuit together form a DC-DC-converter of the type up-converter.

The operation of the circuit arrangement shown in FIG. 1 is as follows.

When circuit input terminals 4 and 5 are connected to a low frequencysupply voltage source, a DC voltage with the approximate shape of arectified sine wave is present between diode bridge output terminals 8and 9. Circuit part 13 renders the switching element 12 alternatelyconductive and non-conductive at a high frequency. As a result, aDC-voltage with an amplitude higher than the maximum amplitude of thelow frequency supply voltage is present across capacitor 15 supplyingthe load. The current through the inductive element increases linearly,when the switching element 12 is conductive and decreases linearly whenthe switching element 12 is non-conductive. The up-converter is operatedin the critical mode. This means that the switching element is renderedconductive when the current through the inductive element has becomeequal to zero. As a result the shape of the current through theinductive element is a series of triangles. Theoretically the averagecurrent drawn from the low frequency supply voltage source has a shapeidentical to and in phase with the low frequency supply voltage, in casethe conduction time of the switching element is constant over each halfa period of the low frequency supply voltage. In practice however,because of for instance filter capacitances or parasytic capacitances,it is found that no current is drawn in the vicinity of the zerocrossings of the low frequency supply voltage causing a substantialamount of THD. Such a parasytic capacitance or capacitor may forinstance be present between circuit input terminals 4 and 5.Alternatively such a parasytic capacitance or capacitor may be presentbetween diode bridge output terminals 8 and 9. These parasyticcapacitances or capacitors are not shown in FIG. 1. In a circuitarrangement according to the present invention the control signal is thesum of a first signal S1, that is substantially constant over each halfa period of the low frequency supply voltage and a second signal S2,that is periodical and has the same phase and frequency as the rectifiedlow frequency supply voltage. As a result, the control signal effects amodulation of the conduction time of the switching element 12. Thismodulation is chosen so that the conduction time of the switchingelement is increased in the vicinity of the zero crossings of the lowfrequency supply voltage. Consequently, more current is drawn from thelow frequency supply voltage source in the vicinity of the zerocrossings of the low frequency supply voltage so that the power factoris increased and the THD is lowered.

FIGS. 2-7 show an implementation of circuit part I and differentimplementations of circuit part II. In each of the FIGS. 2-7 the shapeof the control signal that is the sum of signals S1 and S2 generated bycircuit part I and circuit part II respectively is shown.

In the circuitry shown in each of the figures a circuit part OTA forminga transconductance amplifier is present is present. A transconductanceamplifier is a circuit part that generates an output current that isproportional to the difference between the voltages present at its inputterminals. A first input terminal of transconductance amplifier OTA isconnected to output terminal 16 and a second input terminal is connectedto the output terminal of a circuit part Voutref. The circuit partVoutref generates a signal that represents a desired value of the outputvoltage of the up-converter. An output terminal of the transconductanceamplifier is connected to a terminal Tcontrol. In FIG. 2, a capacitorZcomp connects the terminal Tcontrol to diode bridge output terminal 9.The circuit parts OTA and Voutref together with capacitor Zcomp form acircuit part I for for generating a first signal S1 exclusivelydepending on a desired value and the actual value of the voltage betweenthe output terminals. The value of signal S1 is substantially constantduring each half period of the low frequency supply voltage. It isremarked that the signal representing a desired value of the outputvoltage of the up-converter can be adjustable by a user. The terminalTcontrol is connected to diode bridge output terminal 8 by means ofcapacitor Zthd. Capacitor Zthd together with diode bridge 7 form acircuit part II for generating a second signal S2 that is periodical andhas the same phase and frequency as the rectified low frequency supplyvoltage and for generating the control signal by superimposing thesecond signal S2 on the first signal S1. The second signal S2 isdirectly proportional to the rectified low frequency supply voltage. Thecontrol signal present at terminal Tcontrol thus has a higher value whenthe amplitude of the low frequency supply voltage is high than in thevicinity of the zero crossings of the low frequency supply voltage. Thecontrol circuit is so designed that an increase of the control signalcorresponds to a decrease of the conduction time and vice versa Thecontrol signal Vcontrol is present at terminal T control and is the sumof a substantially constant voltage and a voltage that is proportionalto the rectified low frequency supply voltage and is shown in FIG. 2.

In the circuitry shown in FIG. 3 the capacitor Zcomp present in thecircuitry shown in FIG. 2 has been replaced by a parallel arrangement ofa series arrangement of ohmic resistor Rcomp and capacitor Cbig andcapacitor Csmall. The voltage at the output terminal Tcontrol canthereby be more stably controlled at a certain level for instance whenthe power consumed at the output terminals is changed. The functioningof the circuitry shown in FIG. 3 is very similar to that of thecircuitry shown in FIG. 2 and will therefor not be described separately.The shape of the control signal Vcontrol is similar to the controlsignal shown in FIG. 2.

In FIG. 4 the signal S2 is superimposed on the output signal S1 of thetransductance amplifier OTA making use of a series arrangement of twoohmic resistors Rthd1 and Rthd2 instead of making use of capacitor Zthdas in FIG. 2. Ohmic resistor Rthd2 is in series with Zcomp. In FIG. 4the ohmic resistors Rthd1 and Rthd2 together with diode bridge 7 formthe circuit part II.

In FIG. 5 the signal S2 is superimposed on the output signal S1 of theintegrator INT making use of a series arrangement of two ohmic resistorsRcomp and Rthd1 instead of making use of capacitor Zthd as in FIG. 3. InFIG. 5 the ohmic resistors Rthd1 and Rcomp together with diode bridge 7form the circuit part II. The circuit part I is formed by the circuitparts Voutref and OTA, capacitors Cbig and Csmall and ohmic resistorRcomp. The shape of the control signal Vcontrol generated by theembodiments of the control circuit shown in FIG. 4 and FIG. 5 is similarto the shape shown in FIG. 2 or FIG. 3.

In each of the FIG. 2-5 the second signal S2 is directly proportional tothe rectified low frequency supply voltage. As a result, the conductiontime of the switching element is increased in the vicinity of the zerocrossings of the low frequency supply voltage and decreased when theamplitude of the rectified low frequency supply voltage is comparativelyhigh. In the circuitry shown in FIGS. 6 and 7 the shape of the secondsignal S2 is different.

The circuitry shown in FIG. 6 can be thought of as derived from thecircuitry in FIG. 4 by the addition of a zener diode Zthd shunting ohmicresistor Rthd2. When the amplitude of the rectified low frequency supplyvoltage is low, the zener diode Zthd is non-conductive and the signal S2is therefor proportional to the rectified low frequency supply voltage.When the amplitude of the rectified low frequency supply voltageincreases, the zener diode becomes conductive and the amplitude ofsignal S2 therefor is substantially constant. The resulting shape of thecontrol signal is also shown in FIG. 6. As a consequence the signal S2decreases the amplitude of the control signal and thereby increases theconduction time of the switching element only in the vicinity of thezero crossings of the low frequency supply voltage. In the circuitryshown in FIG. 6, ohmic resistors Rthd1 and Rthd2, zener diode Zthd anddiode bridge 7 together form the circuit part II for generating signalS2.

The circuitry shown in FIG. 7 can be thought of as derived from thecircuitry in FIG. 3 by placing a zener diode Zthd in series with ohmicresistor Rthd1 and shunting Rthd2 with a diode Dthd. In the directvicinity of the zero crossings of the low frequency supply voltage theamplitude of the rectified low frequency supply voltage is so low thatthe zener diode Zthd is non-conductive and signal S2 equals zero. Whenthe momentary amplitude of the rectified low frequency supply voltagebecomes is high enough to render zener diode Zthd conductive, the signalS2 is proportional to the rectified low frequency supply voltagediminished by the voltage drop over zener diode Zthd. When the momentaryamplitude of the rectified low frequency supply voltage increasesfurther, diode Dthd becomes conductive as well. As long as the diodeDthd is conductive, signal S2 increases and decreases far less rapidlythan the rectified low frequency supply voltage, resulting in the shapeof the control signal Vcontrol shown in FIG. 7. Since the modulationdepth is decreased with respect to the embodiments shown in the otherFigures the range over which the conduction time is varied by themodulation is smaller allowing a simpler design of the control circuit.

In a practical embodiment of the circuit arrangement shown in FIG. 1supplied with a low frequency supply voltage with an rms value of 230Volt and a frequency of 50 Hz, it was found that the THD was 14.6% incase the signal S2 was zero, in other words when no modulation of theconduction time of the switching element was effected. In case a controlcircuit as shown in FIG. 2 was used, the THD was reduced to 10.1% and incase a control circuit as shown in FIG. 6 was used the THD was reducedto 6%.

1. A circuit arrangement comprising circuit input terminals forconnection to a low frequency supply voltage source, circuit outputterminals, a rectifier coupled to the circuit input terminals andequipped with rectifier output terminals, a DC-DC-converter coupledbetween the rectifier output terminals and the circuit output terminalsand comprising an inductive element, a unidirectional element coupled tothe inductive element, a switching element coupled to the inductiveelement and the unidirectional element, a control circuit coupled to acontrol electrode of the switching element for rendering the switchingelement alternately conductive and non-conductive at a high frequencyand for generating a control signal for controlling the conduction timeof the switching element in each high frequency period, said controlcircuit comprising a circuit part I for generating a first signal S1that is either a constant or exclusively depends on the actual voltagebetween the output terminals and a desired value of the voltage betweenthe output terminals, characterized in that the control circuit isfurther equipped with a circuit part II for generating a second signalS2 that is periodical and has the same phase and frequency as therectified low frequency supply voltage and for generating the controlsignal by superimposing the second signal S2 on the first signal S1. 2.A circuit arrangement according to claim 1, wherein the second signal S2is proportional to the rectified low frequency supply voltage.
 3. Acircuit arrangement according to claim 1, wherein the second signal isproportional to the rectified low frequency supply voltage when theamplitude of the rectified low frequency supply voltage is lower than apredetermined reference value and substantially equal to thepredetermined reference value when the amplitude of the rectified lowfrequency supply voltage is higher than the predetermined referencevalue.
 4. A circuit arrangement according to claim 1, wherein thecircuit part II comprises a coupling capacitor coupled between therectifier and circuit part I.
 5. A circuit arrangement according toclaim 1, wherein the circuit part II comprises a series arrangement of afirst impedance and a second impedance, coupled between the rectifieroutput terminals and wherein a common terminal of the impedances iscoupled to an output of the circuit part I.
 6. A circuit arrangementaccording to claim 5, wherein at least one of the the impedances is avoltage dependant impedance.
 7. A circuit arrangement according to claim6, wherein the first impedance comprises an ohmic resistor and thesecond impedance comprises a parallel arrangement of an ohmic resistorand a zener diode
 8. A circuit arrangement according to claim 6, whereinthe first impedance comprises a series arrangement of a zener diode andan ohmic resistor and the second impedance comprises an ohmic resistor.9. A circuit arrangement according to claim 6, wherein the secondimpedance comprises an ohmic resistor shunted by a diode.
 10. A circuitarrangement according to claim 1, wherein the DC-DC-converter is anup-converter.
 11. A ballast circuit for operating a lamp comprising acircuit arrangement according to claim 1.